Traveling wave power combiner and radio base station

ABSTRACT

A travelling wave power combiner ensures high isolation between power amplifiers and reduces harmonic components caused by non-linearity of each power amplifier. Therefore, the travelling wave power circuit is provided with even-numbered power amplifiers, a plurality of series-connected branch circuits for respectively distributing input power to the even-numbered power amplifiers, and a plurality of series-connected combiners for combining respective output power of the even-numbered power amplifiers. With respect to all of such combinations as to equalize electric lengths of transmission lines for connecting between the power amplifiers to one another, which combinations are established for the plurality of power amplifiers, the electric lengths at which the combinations are established, take λ/2 i  (where λ=the wavelength of a fundamental wave, and i=positive integer).

BACKGROUND OF THE INVENTION

A distribution power combiner of such a type as explained in theMicrowaves & RF vol. 8, pp 107-112 1998 has been known as a powercombiner applied to a power amplifier of a cellular base station. FIG.18 shows a circuit configuration of the conventional distribution powercombiner. A combining number will be defined as n. A RF signal appliedfrom an input terminal 7 is distributed to respective amplifiers 101-1through 101-n by branch circuits 103-1 through 103-(n−1) in sequence.The RF signals amplified by the respective amplifiers 101 aresequentially combined into one by combiners 102-1 through 102-(n−1),after which the combined one is outputted to an output terminal 8.

In order to bring out the full linearity of a semiconductor device usedin each amplifier 101, it is desirable to reduce the maximum powerinputted to each amplifier 101, i.e., equalize respective power inputtedto the respective amplifiers 101. At this time, the power to beamplified by the respective amplifiers 101 become equal to each other.

Power distribution ratios for equalizing the power inputted to therespective amplifiers 101 are represented every branch circuits. Ifpower inputted to an input transmission line 106 from the input terminal7 is defined as n, it is then necessary to distribute power to beinputted to the first amplifier 101-1 and power to be inputted to thesecond branch circuit 103-2 in a ratio of 1:(n−1). Distribution ratiosto other branch circuits are determined in a similar relationship.

On the other hand, if power outputted from the output terminal 8 throughan output transmission line 107 is defined as n, it is then necessary tocombine first power outputted from the nth amplifier 101-n and (n−1)thpower outputted from the n-1th combiner 102-(n−1) into one. Distributionratios to other combiners are determined in a similar relationship.

SUMMARY OF THE INVENTION

Such a prior art is accompanied by a problem that compensation for thelinearity of each amplifier 101 is not taken into consideration and apower component of harmonic components caused by distortion developed ineach amplifier is outputted from the output terminal 8 of the powercombiner. Thus, in the conventional power amplifier, a filter forcutting off these harmonic components must be inevitably connected to astage subsequent to the power combiner. The efficiency of power has beenimpaired as the entire power combiner including the filter due to aninsertion loss produced by the insertion of the filter.

In the aforementioned prior art, the branching and combining of powerhave been performed by directional couplers. As the prior art thatperforms the branching and combining of the power, there is known oneutilizing impedance ratios between lines in addition to the directionalcouplers. The branching and combining of the power are performedaccording to the impedance ratios between the lines. If the first branchcircuit 103-1 is explained by way of example, then the characteristicimpedance of a line extending from the input terminal 7 to a firstconnecting point, the characteristic impedance of a line extending fromthe first connecting point to the first power amplifier, and thecharacteristic impedance of a line extending from the first connectingpoint to the second branch circuit (second connecting point) may bedefined as Z0/n, Z0 and Z0/(n−1) respectively.

However, when the branching and combining of the power are performedaccording to the impedance ratios between the lines, isolation isinsufficient between the plurality of amplifiers 101 while the circuitcan be constructed at low cost as compared with the use of thedirectional couplers. Therefore, there is the potential that since therelationship in impedance between the distribution power amplifiers aredisturbed when characteristic changes occur in any or more of theamplifiers 101, the efficiency of power combining will be greatlyimpaired. There is also a possibility that if one amplifier developstrouble and the input/output impedance thereof becomes infinite, thenthe output impedances of other amplifiers rise and the power reflectedonto each amplifier increases, thus causing trouble that the amplifierswill be destroyed due to the reflected power on a chain reaction basis.

In a travelling wave power combiner, power are not combined into one ifpower outputted from respective amplifiers are not kept in balance. Inthe travelling wave power combiner according to the present invention,harmonics outputted from amplifiers are canceled out while the poweroutputted from the respective amplifiers are being kept in balance.

The principle of the present invention is shown in FIGS. 19A and 19B.The lengths (electric lengths) of transmission lines for connectingbetween branch circuits or between combiners will be defined as L (forsimplification, the branch circuits and combiners are omitted in thedrawing). FIG. 19A shows the relationship between the differences inphase between kth harmonics developed in an output point 1901 and linelengths L for canceling out the phase differences. The wavelength of afundamental wave is defined as λ and a phase shift of φ will bedeveloped due to each line length L. If the phase of an input point 1900is defined as the reference, a kth harmonic developed from a firstamplifier 1902 produces a phase shift of kφ at the output point 1901through a transmission line 1905. On the other hand, a kth harmonicproduced from a second amplifier 1903 produces a phase shift of φ at theoutput point 1901 through a transmission line 1904. Thus, the phasedifference Δφ at the output point 1901 between the kth harmonicsdeveloped in the two amplifiers results in (k−1)φ. When the phase of thekth harmonic from the first amplifier 1902 and the phase of the kthharmonic from the second amplifier 1903 are reversed, their kthharmonics are canceled out. In the case of Δφ=λ/2, the phases of the kthharmonics from the two amplifiers are inverted. Line lengths in thiscase are shown so as to correspond to second to 5th harmonics.

On the other hand, it is necessary to keep the power outputted from therespective amplifiers in balance with a view toward functioning as thetravelling wave power combiner. This means that the combining orcancellation of the power (fundamental waves or harmonics) outputtedfrom the respective amplifiers need to be equally executed for all theamplifiers. If, for example, harmonics from only some of the amplifiersin the travelling wave power combiner are canceled out, there is then adifference in magnitude between fundamental-wave outputs of theharmonics-canceled amplifiers and the amplifiers free of thecancellation of the harmonics, whereby the combiner is kept off-balance.

In order to make combinations of the amplifiers for canceling theharmonics, the number of the amplifiers contained in the travelling wavepower combiner is set to a even number. Examples of 4-combining (1),6-combining (2), 8-combining (3) and 10-combining (4) are shown in FIG.19B. If the 4-combining are taken, for example, then the number ofcombinations of the amplifiers used for cancellation is two. The firstcombination is amplifiers {(1, 2), (3, 4)} and the second combination isamplifiers {(1, 3), (2, 4)}. Thus, a transmission-line length (electriclength) between the amplifiers (1, 2) or a transmission-line lengthbetween the amplifiers (1, 3) may be defined as a transmission-linelength (=λ/(2(k−1)) for canceling the kth harmonic. If thetransmission-line length between the amplifiers (1, 2) is set to λ/8,for example, then a 5th harmonic is canceled out by the amplifiers (1,2) and a 3rd harmonic is canceled out by the amplifiers (1, 3). Inregard to these combinations, the transmission lines for connectingbetween the amplifiers are equal in electric length to each other.

It is further necessary for the travelling wave power combiner includingthe even-numbered amplifiers that a plurality of ways to make thecombinations of the amplifiers for canceling harmonics exist and theharmonics are canceled out for all possible combinations. If even someof the possible combinations cannot cancel out the harmonics, then adifference in magnitude occurs between the outputs of the fundamentalwaves and hence the combiners are kept off-balance. In the exampleillustrated in FIG. 19B, the transmission-line lengths between therespective adjacent amplifiers are set equal to each other. In thiscase, the harmonics may be canceled out in the case of any combinationof other than the adjacent amplifiers.

It is desirable that the combinations of non-adjoining amplifiers aredetermined according to the transmission-line lengths. In this case, theentire circuit can be constructed in compact form. Further, since thepower of the high-order harmonics outputted from the amplifiers isgenerally high in energy as the order of harmonics decreases, it isdesirable to cancel out lower-order harmonics, particularly, the secondand third harmonics. It is practical that as will be described later,the third harmonics can be canceled out in the combinations of thenon-adjoining amplifiers.

Since the harmonics to be canceled out are kept in balance on anamplitude basis and outputted from the respective amplifiers whosecharacteristics are the same, they are not outputted to the outputterminal of the travelling wave power combiner. Thus, in the poweramplifier, the connection of the filter for cutting off the harmoniccomponents to the stage subsequent to the travelling wave power combinerbecomes unnecessary, whereby an impairment in the power efficiency dueto an insertion loss produced by the insertion of the filter is solved.

These and other objects and many of the attendant advantages of theinvention will be readily appreciated as the same becomes betterunderstood by reference to the following detailed description whenconsidered in connection with the accompanying drawing

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram showing a first configuration example of atravelling wave power combiner (first embodiment) according to thepresent invention;

FIG. 2 is a block diagram illustrating a second configuration example ofthe travelling wave power combiner (first embodiment);

FIG. 3 is a block diagram depicting a third configuration example of thetravelling wave power combiner (first embodiment);

FIG. 4 is a block diagram showing a fourth configuration example of thetravelling wave power combiner (first embodiment);

FIG. 5 is a planar circuit pattern diagram corresponding to the firstconfiguration example of the travelling wave power combiner (firstembodiment);

FIG. 6 is a planar circuit pattern diagram corresponding to the secondconfiguration example of the travelling wave power combiner (firstembodiment);

FIG. 7 is a planar circuit pattern diagram corresponding to the thirdconfiguration example of the travelling wave power combiner (firstembodiment);

FIG. 8 is a planar circuit pattern diagram corresponding to the firstconfiguration example of the travelling wave power combiner (firstembodiment);

FIG. 9 is a planar circuit pattern diagram corresponding to the firstconfiguration example of the travelling wave power combiner (firstembodiment);

FIG. 10 is a planar circuit pattern diagram corresponding to the secondconfiguration example of the travelling wave power combiner (firstembodiment);

FIG. 11 is a block diagram showing a first configuration example of atravelling wave power combiner (second embodiment);

FIG. 12 is a block diagram illustrating a second configuration exampleof the travelling wave power combiner (second embodiment);

FIG. 13 is a block diagram depicting a third configuration example ofthe travelling wave power combiner (second embodiment);

FIG. 14 is a block diagram showing a fourth configuration example of thetravelling wave power combiner (second embodiment);

FIG. 15 is a planar circuit pattern diagram corresponding to the firstconfiguration example of the travelling wave power combiner (secondembodiment);

FIG. 16A is a diagram showing a structure of a cellular system;

FIG. 16B is a diagram illustrating a structure of a cellular basestation;

FIG. 17A is a diagram depicting a transmitting/receiving amplifier of acellular base station;

FIG. 17B is a block diagram showing a linearized power amplifier for thecellular base station;

FIG. 18 is a block diagram illustrating a configuration of aconventional distribution power combiner;

FIG. 19A is a first diagram for describing the principle of the presentinvention; and

FIG. 19B is a second diagram for describing the principle of the presentinvention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

A cellular system shown in FIG. 16A is adopted to repeatedly use limitedfrequencies with efficiency and increase the capacity of subscribersheld or accommodated by the system in the mobile communication. In thecellular system, the directivity of an antenna is sharpened to localizean achievable distance of a radio wave transmitted or emitted from onebase station (cells 1601 through 1605), whereby the same frequency isspatially re-used. For example, the same frequency f1 is used betweenthe cells 1601 and 1603. A structure of the base station is shown inFIG. 16B. A signal supplied from a base band signal processing device(not shown) is transformed into a frequency (carrier frequency) capableof propagating through space as a radio wave by modulation demodulationequipment 1610. A transmitting/receiving amplifier 1611 amplifies thepower of a transmitting/receiving signal to allow the radio wave toreach each terminal lying within the corresponding cell (on thetransmitting side) or to extract a weak signal sent from the terminal(on the receiving side). The signal amplified by a transmitting power(linearized) amplifier 1007 is transmitted to each cell as power havinga sharp directivity formed by an array antenna 1612.

In the current cellular mobile communication system, the achievabledistance of a radio wave sent or emitted from a base station is a few kmto about 20 km, the frequency to be used ranges from several hundreds ofMHz to a few GHz, and radiation power of one terminal ranges fromseveral hundreds of mW to about 2 W in terms of a demand for a sizereduction in terminal, the amount of information transmission requiredby the system and space attenuation characteristics of anelectromagnetic wave. In this case, average transmission power, whichranges from several tens of W to several hundreds of W, is required asthe output of the linearized power amplifier 1007 to allow one basestation to support several tens to several hundreds of subscribers.

Further, since the phase/amplitude modulation system is adopted in thedigital mobile communication system which is on the mainstream toimplement a variety of communication services, high linearity isrequired of the linearized power amplifier 1007. In order to make up fornon-linearity of a semiconductor device for implementing the linearizedpower amplifier, the efficiency thereof is generally low and valuesranging from several hundreds of W to a few KW as saturation power arerequired of the linearized power amplifier. Therefore, an improvement inthe efficiency of the linearized power amplifier brings about a greateffect at reducing power consumption of the base station.

The amplification of transmitting power is performed by a linearizedpower amplifier 1007 in a transmitting/receiving amplifier shown in FIG.17A. A received signal is amplified by a low noise amplifier 1006. FIG.17B shows an example of a configuration of a feed forward type poweramplifier as one typical linearized power amplifier 1007. A main signalamplified by a main power amplifier 1030 is divided by a branch circuit1021 and combined with the pre-amplification main signal by a combiner1022. Thus, an error signal (distortion signal) caused by theamplification of the main signal is extracted (by an error signaldetection circuit). The extracted error signal is amplified by an erroramplifier 1031. The amplified error signal is combined with theamplified main signal by the combiner 1023, whereby distortion iseliminated from the amplified main signal (by a distortion eliminationcircuit). A feedforward controller 1032 monitors power and controlsvariable phasers 1013 and 1014 to optimize the elimination ofdistortion.

Since the linearity of the current semiconductor power amplifier usingthe semiconductor is generally insufficient, a large-output main poweramplifier 1030 is built in a linearized power amplifier to therebyensure the linearity of power amplification required of the system. Asemiconductor device used for the power amplifier is normally low inone-device outputtable power and provides a saturation output whichranges from about 10 W to 200 W at a 2 GHz band under the presenttechnical level, for example. Thus, the main power amplifier 1030 makesuse of a plurality of amplifiers each comprised of a semiconductordevice and merges or combines their outputs into one.

FIG. 1 shows a first configuration example of a travelling wave powercombiner used as the main power amplifier 1030. A combining number willbe defined as n. Directional couplers are applied to branch circuits 1and combiners 2, and transmission lines 4 a and 4 b are provided. Aninput terminal of a branch circuit 1-1 is coupled to an input terminal 7through an input transmission line 5. A first output terminal of thebranch circuit 1-1 is electrically connected to a power amplifier 3-1,whereas a second output terminal thereof is electrically coupled to aninput terminal of a branch circuit 1-2 through a coupling transmissionline 4 b-1. The output of the power amplifier 3-1 is coupled to a firstinput terminal of a combiner 2-1 through the coupling transmission line4 a-1. The output of a power amplifier 3-2 is coupled to a second inputterminal of the combiner 2-1 and an output terminal thereof is coupledto a first input terminal of a combiner 2-2 through a couplingtransmission line 4 a-2.

Similarly, a second output terminal of a branch circuit 1-(i−1) iscoupled to an input terminal of a branch circuit 1-i (2≦i≦n−2) through acoupling transmission line 4 b-(i−1). A first output terminal of thebranch circuit 1-i is electrically connected to a power amplifier 3-iand a second output terminal thereof is electrically connected to aninput terminal of a branch circuit 1-(i+1) through a couplingtransmission line 4 b-i.

Further, an output terminal of a combiner 2-(i−1) is coupled to a firstinput terminal of a combiner 2-i (2≦i≦n−2) through a couplingtransmission line 4 a-i, whereas the output of a power amplifier 3-(i+1)is coupled to a second input terminal thereof. An output terminalthereof is coupled to a first input terminal of a combiner 2-(i+1)through a coupling transmission line 4 a-(i+1).

A branch circuit 1-(n−1) and a combiner 2-(n−1) have connectionrelations similar to the above. However, a second output terminal of thebranch circuit 1-(n−1) is coupled to a power amplifier 3-n through acoupling transmission line 4 b-(n−1). Further, an output terminal of thecombiner 2-(n−2) is coupled to a first input terminal of the combiner2-(n−1) through a coupling transmission line 4 a-(n-1), whereas a secondinput terminal thereof is coupled to the output of the power amplifier3-n. The output of the combiner 2-(n−1) is coupled to an output terminal8 through an output transmission line 6. Incidentally, branching ratios(combining ratios) of branch circuits (combiners) are shown in thedrawing.

When the directional couplers are used for the branch circuits 1 and thecombiners 2, isolation between the respective power amplifiers 3 can bekept high. In the directional coupler, the amount of coupling thereof isdetermined according to an interval of a parallel conductor whichconstitutes a coupler, and each coupling transmission line 4, inputtransmission line 5 and output transmission line 6 can be set equal inimpedance. When these lines are formed of a micro strip line, the widththereof can be kept constant, so that the area of the power combiner canbe restrained.

In the present invention, an electric length L of the couplingtransmission line 4 will be defined based on the above-describedprinciple. An electric length between power amplifiers utilized incombination in the travelling wave power combiner may be set so as totake λ/(2i) (where λ=wavelength of fundamental wave and i=positiveinteger) and transmission lines between the adjacent power amplifiersmay be set equal in length to each other. At this time, the power of an(i+1)th harmonic outputted from a power amplifier separated by λ/(2i) isreturned to the power amplifier again, where it is converted to afundamental wave component. It is thus possible to improve theefficiency of combining of outputs from the power amplifiers.

FIG. 2 is a block diagram showing a second configuration example of thetravelling wave power combiner (first embodiment). The travelling wavepower combiner has a feature in that the branch circuit 1-(n−1) and thecombiner 2-1 shown in the configuration example of FIG. 1 arerespectively replaced with a wilkinson-type power branch circuit 9 and awilkinson-type power combiner 10. A branching ratio and combining ratioof a branch circuit 1-n and a combiner 2-1 are 1:1 respectively. Whenthe branch circuit 1-n and the combiner 2-1 are respectively constructedof a directional coupler having a monolayered structure usingelectromagnetic coupling, the required amount of electromagneticcoupling increases and the fabrication of its construction falls intodifficulties. The wilkinson-type power branch circuit (combiner) havingthe branching ratio (combining ratio) of 1:1 provides the effects ofbeing capable of being easily manufactured by a planar circuit having amonolayered structure and reducing its cost.

FIG. 3 is a block diagram showing a third configuration example of thetravelling wave power combiner (first embodiment). A first feature ofthe present configurational example resides in that short stubs 52 areconnected to their corresponding output terminals of power amplifiers 3.Described specifically, combiners 2 are implemented by means of quarterdirectional couplers and each short stub 52 has an electric lengthcorresponding to a wavelength equal to one-fourth of a fundamental wave(f). Since the output terminals of the power amplifiers areshort-circuited as to even-order harmonics (2 f, 4 f . . . ) owing tothe provision of such short stubs, even-order harmonic power is fed backto the power amplifiers and some thereof is converted to a fundamentalwave component. Therefore, the efficiency of the travelling wave powercombiner increases.

It is desirable that when such a construction is utilized, thecomponents of the even-order harmonics are canceled out each other bythe short stubs and the components of odd-order harmonics are canceledout each other according to the selection of transmission line lengths.A third harmonic and a 5th harmonic can be canceled out by takingm={fraction (1/8+L )} in the case of the 8-combining, for example. Sincethe distortion of power outputted from each combiner is improved wherethe travelling wave power combiner is constructed so as to cancel outthe odd-order harmonics each other, the operating point can be increasedand hence the efficiency of the combiner can be improved.

A second feature resides in that open stubs 54 each having an electriclength corresponding to a wavelength equal to one-twelfth thefundamental wave and resonant inductors 53 whose one ends are grounded,are coupled in parallel to matching terminators of the combiners 2. Thevalue of inductance of the resonant inductor 53 is selected so as toresonate in parallel with the open stub 54 at a frequency of thefundamental wave. For example, the inductance value thereof results inabout 6 nH at a fundamental wave 2 GHz (frequency employed in thecellular system). Thus, this can be implemented with a practicallysufficient Q value. Since, at this time, the matching terminators of the¼-wavelength directional couplers (combiners 2) are shorted as to thethird harmonic (3f), third harmonic power is fed back to the poweramplifiers without being consumed through the resistance of the matchingterminator of each combiner 2 and some thereof is converted to afundamental wave component. Therefore, the efficiency of the travellingwave power combiner can be kept high even when the respective poweramplifiers are out of balance.

FIG. 4 is a block diagram showing a fourth configuration example of thetravelling wave power combiner (first embodiment). The branch circuit1-(n−1) and combiner 2-1 shown in the configuration example of FIG. 3are replaced with a wilkinson-type power branch circuit 9 and awilkinson-type power combiner 10 respectively.

FIG. 5 is a pattern diagram corresponding to the configuration exampleshown in FIG. 1 implemented by planar circuits. The drawing shows aplanar pattern of the travelling wave power combiner at n=4. An inputterminal of a branching microstrip directional coupler 11-1 (branchcircuit 1) is coupled to an input terminal 7 via an input microstripline 15 (input transmission line 5). A first output terminal of thebranching microstrip directional coupler 11-1 is coupled to a surfacemount type power amplifier 13-1 (power amplifier 3), whereas a secondoutput terminal thereof is coupled to a first input terminal of abranching microstrip directional coupler 11-2 through a couplingmicrostrip line 14 b-1 (coupling transmission line 4 b). Incidentally,an input matching microstrip line 20 b-1 and an output matchingmicrostrip line 20 a-1 are respectively connected to input/outputterminals of the surface mount type power amplifier 13-1. The output ofthe surface mount type amplifier 13-1 is coupled to a first input of acombining microstrip directional coupler 13-1 (combiner 2) through acoupling microstrip line 14 a-1 (coupling transmission line 4 a).

Further, an output terminal of a combining microstrip directionalcoupler 12-2 is coupled to a first input terminal of a combiningmicrostrip directional coupler 12-3 through a coupling transmission line14 a-3, and a surface mount type power amplifier 13-4 is coupled to asecond output terminal thereof. An output terminal of the combiningmicrostrip directional coupler 12-3 is coupled to an output terminal 8through an output microstrip line 16 (output transmission line 6).Incidentally, those shown in the parentheses indicate theircorresponding circuit configurations of FIG. 1. Further, each of thebranching microstrip directional coupler 11 and combining microstripdirectional coupler 12 makes use of a {fraction (1/4+L )}-wavelengthmicrostrip directional coupler, for example.

The following characteristics are included in the planar pattern shownin FIG. 5. The first is that the longitudinally-extending central axesof linear gaps between microstrip couplers each constituting thebranching microstrip directional coupler are placed linearly withrespect to the respective branching microstrip directional couplers 11-1through 11-3. The combining microstrip directional coupler 12 is alsosimilar to the above. The directions of equivalent magnetic currentsources developed in the linear gaps are radiating directions null eachother at both ends of the microstrip couplers each constituting themicrostrip directional coupler. Therefore, electromagnetic couplingbetween the branching (combining) microstrip directional couplers 11(12) is restrained so that the operation of the travelling wave powercombiner is stabilized.

The second is that the linear gap between the adjacent branchingmicrostrip directional couplers 11 and the linear gap between theadjacent combining microstrip directional couplers 12 are not opposed toeach other or the opposite ones are provided so as to be as small aspossible. In the example shown in FIG. 5, the linear gaps between thebranching microstrip directional couplers 11-1 through 11-3 arerespectively provided so as to be opposed to the coupling microstriplines 14 a-1 through 14 a-3, whereas the linear gaps between thecombining microstrip directional couplers 12-1 through 12-3 arerespectively provided so as to be opposite to the coupling microstriplines 14 b-1 through 14 b-3. If the linear gap between the adjacentbranching microstrip directional couplers 11 and the linear gap betweenthe adjacent combining microstrip directional couplers 12 are placed inopposing relationship, then each microstrip coupler of the branchingmicrostrip directional coupler 11 and each microstrip coupler of thecombining microstrip directional coupler 12 are electromagneticallycoupled therebetween. Thus, this might exert a bad influence on thecircuit operation. The operation of the travelling wave power combineris stabilized by placing the equivalent magnetic current sourcesdeveloped in the linear gaps between the branching microstripdirectional couplers and the equivalent magnetic current sourcesdeveloped in the linear gaps between the combining microstrip couplersso as not to be directly opposed to one another.

FIG. 6 is a pattern diagram corresponding to the configuration exampleshown in FIG. 2 implemented by planar circuits. The drawing shows a flator planar pattern of the travelling wave power combiner at n=4. Ifcompared with the planar pattern shown in FIG. 5, then the branchingmicrostrip directional coupler 11-3 is replaced by a planarwilkinson-type two branch divider or circuit 21 (wilkinson-type powerbranch circuit 9) and the combining microstrip directional coupler 12-1is replaced with a planar wilkinson-type two branch divider or circuit22 (wilkinson-type power branch circuit 10), respectively. As describedin regard to FIG. 2, the accuracy of fabrication of the planar circuitpattern can be lessened and the manufacturing cost can be reduced.Incidentally, those shown in the parentheses indicate theircorresponding circuit configurations of FIG. 2.

FIG. 7 is a pattern diagram corresponding to the configuration exampleshown in FIG. 3 implemented by planar circuits. The drawing shows aplanar pattern of the travelling wave power combiner at n=4. Shortmicrostrip stubs 62 (short stubs 52) each having an electric lengthcorresponding to a wavelength equal to one-fourth the fundamental waveare respectively connected to output terminals of respective surfacemount type power amplifiers 13. Open microstrip stubs 63 (open stubs 54)each having an electric length corresponding to a wavelength equal toone-twelfth the fundamental wave, and resonant chip inductors 64(resonant inductors 53) whose one ends are grounded, are respectivelyconnected in parallel with matching terminators of respective combiningmicrostrip directional couplers 12. Incidentally, those shown in theparentheses indicate their corresponding circuit configurations of FIG.3.

FIG. 8 shows a first modification of the planar pattern shown in FIG. 5.Longitudinal directions of respective coupling microstrip lines 14 a (14b) are placed so as to take directions substantially orthogonal tolongitudinal directions of combining microstrip directional couplers 12(respective branching microstrip lines 11). Owing to such placement, theinterval between an input terminal 7 and an output terminal 8 of thetravelling wave power combiner can be shortened and the area thereof canbe reduced.

A combining microstrip directional coupler 12-1 and a branchingmicrostrip directional coupler 11-3 may be replaced by a planarwilkinson-type double combiner and a planar wilkinson-type two branchdividers or circuits, respectively (corresponding to the configurationexample shown in FIG. 2). The planar pattern shown in FIG. 8 may ofcourse be applied even to the configuration examples shown in FIGS. 3and 4.

FIG. 9 shows a second modification of the planar pattern shown in FIG.5. The branching microstrip directional couplers 11 and the combiningmicrostrip directional couplers 12 are replaced with {fraction (1/4+L)}-arcuate branching wavelength microstrip directional couplers 31 and{fraction (1/4+L )}-arcuate combining microstrip directional couplers 32respectively. In the structure shown in FIG. 9, the respective {fraction(1/4+L )}-arcuate branching microstrip directional couplers (respective{fraction (1/4+L )}-arcuate combining microstrip directional couplers)are placed in a parallel translation relationship to one another.

The travelling wave power combiner is stably operated because equivalentmagnetic current sources developed in linear gaps between the {fraction(1/4+L )}-arcuate branching microstrip directional couplers andequivalent magnetic current sources developed in linear gaps between the{fraction (1/4+L )}-arcuate combining microstrip directional couplersare not directly opposed to one another due to the formation of thelinear gaps between the microstrip directional couplers in arcuate form.A transverse size extending from the input terminal 7 to the outputterminal 8 is also shortened to substantially two-third that employed inthe travelling wave power combiner shown in FIG. 5.

As shown in FIG. 10, a {fraction (1/4+L )}-arcuate combining microstripdirectional coupler 32-1 and a {fraction (1/4+L )}-arcuate branchingmicrostrip directional coupler 31-3 may be replaced by a planarwilkinson-type double combiner 22 and a planar wilkinson-type two branchdivider or circuit 21 respectively in the planar pattern shown in FIG. 9(corresponding to the configuration example shown in FIG. 2). The planarpatterns shown in FIGS. 9 and 10 can be of course applied even to theconfiguration examples shown in FIGS. 3 and 4.

FIG. 11 shows a first configuration example of a second embodiment of atravelling wave power combiner used as a main power amplifier 1030. Acombining number will be defined as n. In the second embodiment, piecesof power are distributed or combined into one according to an impedanceratio between lines. A signal inputted to an input terminal 111 isdistributed by means of a first branch circuit (corresponding to aninput matching transmission line 101 f 1 and a branching transmissionline 101 a 1) through an input transmission line 101 a 0. A first output(corresponding to the output of the input matching transmission line 101f) of the first branch circuit is inputted to a first power amplifier101 c 1, whereas a second output (corresponding to the output of thebranching transmission line 101 a) is inputted to a second branchcircuit (corresponding to 101 f 2 and 101 a 2). Powerbranching/combining ratios are respectively designated at points wherethe input matching transmission lines 101 f and the branchingtransmission lines 101 a are connected to one another. If power inputtedto the input transmission line 101 a 1 from the input terminal 111 isdefined as n, for example, it is then necessary to distribute powerinputted to the first input matching transmission line 101 f 1 and powerinputted to the first branching transmission line 101 a 1 to 1:(n−1).For the purpose of coping with it, the characteristic impedance of theinput transmission line 101 a 0 and the characteristic impedance of thefirst branching transmission line 101 a 1 may be defined as Z0/n andZ0/(n−1) respectively if the characteristic impedance of the inputmatching transmission line 101 f 1 is set as Z0. The impedances of otherinput matching transmission lines and branching transmission lines aredetermined in the same relationship to the above.

The output of the first power amplifier 101 c 1 results in a first input(corresponding to the input of a combining transmission line 101 b) of an-1th combiner (corresponding to a combining transmission line 101b(n−1) and an output matching transmission line 101 e(n−1)) through aline 101 en. Incidentally, a second input (corresponding to the input ofan output matching transmission line 101 e) results in the output of asecond power amplifier 101 c 2.

Similarly, a second output of an i−1th branch circuit is inputted to anith branch circuit (2≦i≦n−2). A first output (101 f) thereof is inputtedto a power amplifier 101 ci. A second output (101 a) is inputted to ani+1 th branch circuit. The output of an n-(i−1) the combiner is inputtedas a first input (101 b side) of an n-ith combiner (2≦i≦n−2), and theoutput of an i+1th power amplifier is inputted as a second input (101 eside).

An n-1 th branch circuit (101 f(n−1), 101 a(n−1)) and a first combiner(101 b 1 and 101 e1) have connection relationships similar to the above.A second output (101 a side) of the n−1th branch circuit is inputted toan nth power amplifier 101 cn through a line 101 fn. The output of thefirst combiner is coupled to an output terminal 112 through an outputtransmission line 101 b 0. Branching ratios or combining ratios andimpedance ratios between the respective lines are represented as shownin the drawing.

As a first feature, electric lengths of the transmission lines 101 a and101 b are determined in a manner similar to the first embodiment. It isthus possible to improve the output combining efficiency of each poweramplifier.

As a second feature, the input side of the power amplifier 101 ci andthe input side of a power amplifier 101 ci+1 (1≦i≦n−1) are respectivelyconnected to one another by a transmission line 101 di for isolation,which includes a series resistor. Here, the transmission line 101 d forisolation has an electric length equal to that of each of thetransmission lines 101 a and 101 b. Owing to such a structure, even ifthe inputs to the power amplifiers are unbalanced, such unbalance can beaccommodated by the series resistors of the transmission lines forisolation, whereby the isolation between the respective power amplifierscan be kept high.

FIG. 12 is a block diagram showing a second configuration example of thetravelling wave power combiner (second embodiment). In addition to theconfiguration shown in FIG. 11, the output side of the power amplifier101 ci and the power amplifier 101 ci+1 (1≦i≦n−1) are connected to eachother by the transmission line 101 di for isolation, which includes theseries resistor. Since the isolation between the outputs of the poweramplifiers is improved in such a construction, it is possible to lessenthe influence of characteristic deviations or the like developed betweenthe respective power amplifiers 101 c as compared with the configurationexample shown in FIG. 11.

FIG. 13 is a block diagram showing a third configuration example of thetravelling wave power combiner (second embodiment). In the presentembodiment as distinct from the embodiment shown in FIG. 11, directionalcouplers are adopted as branch circuits. It is desirable that since apre-amplification displacement or deviation is amplified by a poweramplifier and exerts a great influence on the circuit operation, theisolation is rendered high on the input side.

Directional couplers 108 e 1 through 108 e(n−1) are respectivelyinserted in front of coupling transmission lines 108 a 1 through 108a(n−1) on the input side, which connect between respective poweramplifiers. Power is distributed to the respective power amplifiers bythese directional couplers. As to the degree of coupling of eachdirectional coupler, the degree of coupling of the first directionalcoupler 108 e 1 is 1/n, that of the second directional coupler 108 e 2is 1/(n−1), and that of the n−1th directional coupler 108 e(n−1) is{fraction (1/2+L )}. In the example illustrated in the drawing,{fraction (1/4+L )}-wavelength directional couplers are used as thesedirectional couplers. In order to compensate for phase differencesdeveloped due to the use of the {fraction (1/4+L )}-wavelengthdirectional couplers, phase-compensating transmission lines 108 h 1through 108 h(n−1) each having an electric length corresponding to asubstantially {fraction (1/4+L )} wavelength are provided on the outputside. Further, an input-side phase compensating transmission line 108 jhaving an electric length corresponding to an approximate {fraction(1/2+L )} wavelength is provided posterior to the coupling transmissionline 108 a(n−1) on the input side. A wilkinson-type power branch circuitmay be used as the n−1th directional coupler.

Owing to such a construction, the isolation on the input side of eachpower amplifier can be made high and abnormal oscillations or the likedeveloped due to the influence of input impedance between the poweramplifiers can be lessened. Further, the impedance of each couplingtransmission line 108 a on the input side can be kept constant. It istherefore possible to restrain an increase in circuit scale.

FIG. 14 is a block diagram showing a fourth configuration example of thetravelling wave power combiner (second embodiment). The presentembodiment is different from the embodiment shown in FIG. 13 in thatline lengths of transmission lines 109 f 1 through 109 fn provided onthe output sides of respective power amplifiers are respectively setapproximately to a {fraction (1/4+L )} wavelength and a resistor 109d(i−1) for isolation is provided between an output terminal of an ithpower amplifier 109 ci (2≦i≦n) and the output side of an i−1th couplingtransmission line 109 b(i−1). In a manner similar to the configurationshown in FIG. 12 owing to such a configuration, each power amplifier canbe rendered high in isolation on the output side thereof and theinfluence of characteristic deviations developed between the poweramplifiers can be lessened.

FIG. 15 is a planar pattern showing the configuration of the travellingwave power combiner shown in FIG. 11, which is implemented by a planarcircuit having a monolayered structure. The drawing shows an example ofn=4. A signal is inputted to an input terminal 1101 so as to be by-powerdistributed to an input-side line 10 f 1 of a first transistor chip(power amplifier) 10 c 1 and a coupling transmission line 10 a 1 throughan input microstrip line 10 a 0. The output side also functions in amanner similar to the above. That is, the sum of signals amplified bythe first to third transistor chips passes through a couplingtransmission line 10 b 1, followed by combining with a signal outputtedfrom an output-side line 10 d 4 and amplified by a fourth transistorchip 10 c 4. Thereafter, the combined signal is outputted from an outputterminal 1102 through an output microstrip line 10 b 0. Here, lines 10 e1 through 10 e 3 for isolation, which include isolation resistorsrespectively, are provided between the input-side lines 10 f of therespective transistor chips.

According to the present invention, since harmonic components can becanceled out by a power combiner, each power amplifier can be improvedin power efficiency.

It is further understood by those skilled in the art that the foregoingdescription is a preferred embodiment of the disclosed device and thatvarious changes and modifications may be made in the invention withoutdeparting from the spirit and scope thereof.

What is claimed is:
 1. A power combiner, comprising: even-numbered poweramplifiers; a plurality of series-connected branch circuits forrespectively distributing input power to said even-numbered poweramplifiers; a plurality of series-connected combiners for combiningrespective output power of said even-numbered power amplifiers; openstubs respectively having electric lengths of λ/12 (where λ=wavelengthof a fundamental wave), which are provided at matching terminators ofsaid plurality of combiners; and inductors coupled in parallel with saidopen stubs respectively; wherein said even-numbered power amplifiers areconnected by transmission lines so that in the case of all combinationsin which each of said combinations consists of a plurality of pairs ofsaid power amplifiers with electric lengths of said transmission linesenabling connection between said power amplifiers of each pair of saidplurality of pairs of said power amplifiers to one another beingequalized, said electric lengths at which said combinations areestablished, take λ/2i (where i=positive integer).
 2. The power combineraccording to claim 1, wherein the electric lengths at which said all thecombinations are established, include electric lengths other than theshortest electric length, which are λ/4.
 3. A power combiner,comprising: even-numbered power amplifiers; input matching transmissionlines respectively connected to input terminals of said even-numberedpower amplifiers; output matching transmission lines respectivelyconnected to output terminals of said even-numbered power amplifiers; aplurality of input transmission lines respectively having identicalelectric lengths λ/2i (where i=positive integer), for connecting saidinput matching transmission lines in parallel; and a plurality of outputtransmission lines respectively having electric lengths equal to thoseof said input transmission lines, for connecting said output matchingtransmission lines in parallel; wherein transmission lines forisolation, which have electric lengths equal to said input transmissionlines, and resistors connected in series with said transmission linesfor isolation are respectively provided at input terminals of said poweramplifiers so as to be connected between the adjacent input transmissionlines of said input transmission lines connected in parallel.
 4. Thepower combiner according to claim 3, wherein transmission lines forisolation, which have electric lengths equal to said output transmissionlines, and resistors series-connected to said transmission lines forisolation are provided at output terminals of said power amplifiers soas to be connected between the adjacent output transmission lines ofsaid output transmission lines connected in parallel.
 5. The powercombiner according to claim 3, wherein input power is distributedaccording to impedance ratios between said input matching transmissionlines and said input transmission lines connected to said input matchingtransmission lines, and output power are combined into one according toimpedance ratios between said output matching transmission lines andsaid output transmission lines connected to said output matchingtransmission lines.